No-load-modulation, high-efficiency power amplifier

ABSTRACT

Apparatus and methods for a multiclass, broadband, no-load-modulation power amplifier are described. The power amplifier ( 500 ) may include a main amplifier ( 532 ) operating in a first amplification class and a plurality of peaking amplifiers ( 536, 537, 538 ) operating in a second amplification class. The main amplifier ( 532 ) and peaking amplifiers ( 536, 537, 538 ) may operate in parallel on portions of signals derived from an input signal to be amplified. The main amplifier ( 532 ) may see no modulation of its load impedance between a fully-on state of the power amplifier (all amplifiers amplifying) and a fully backed-off state (peaking amplifiers idle). By avoiding load modulation, the power amplifier ( 500 ) can exhibit improved bandwidth and efficiency compared to conventional Doherty amplifiers.

RELATED APPLICATION

This Application is a national stage filing under 35 U.S.C. 371 ofInternational Patent Application No. PCT/IB2017/001575, filed Oct. 2,2017, entitled “NO-LOAD-MODULATION, HIGH-EFFICIENCY POWER AMPLIFIER,”the entire contents of which is herein incorporated by reference in itsentirety.

BACKGROUND Technical Field

The technology relates to high-speed, high-power amplifiers which may beconstructed from transistors formed from semiconductor materials suchas, but not limited to, gallium nitride.

Discussion of the Related Art

High-speed power amplifiers formed from semiconductor materials have avariety of useful applications, such as radio-frequency (RF)communications, radar, RF energy, power conversion, and microwaveapplications. Gallium nitride semiconductor material has receivedappreciable attention in recent years because of its desirableelectronic and electro-optical properties. Because of its wide bandgap,GaN is more resistant to avalanche breakdown and can maintain electricalperformance at higher temperatures than other semiconductors, such assilicon. GaN also has a higher carrier saturation velocity and cansustain higher power densities compared to silicon. Additionally, GaNhas a Wurtzite crystal structure, is a very stable and hard material,has a high thermal conductivity, and has a much higher melting pointthan other conventional semiconductors such as silicon, germanium, andgallium arsenide. Accordingly, GaN is useful for high-speed,high-voltage, and high-power applications.

Applications supporting mobile communications and wireless internetaccess under current and proposed communication standards, such asWiMax, 4G, and 5G, can place austere performance demands on high-speedamplifiers constructed from semiconductor transistors. The amplifiersmay need to meet performance specifications related to output power,signal linearity, signal gain, bandwidth, and efficiency.

SUMMARY

Apparatus and methods for amplifying radio-frequency signals aredescribed. A multiclass, no-load-modulation power amplifier may includea plurality of amplifiers operating in parallel on portions of areceived signal in different amplification classes and providingamplified signals to a common output of the power amplifier. A mainamplifier may amplify signals at low signal levels and high signallevels. Secondary amplifiers (referred to as peaking amplifiers) may beidle (non-amplifying) at low signal levels and become active(amplifying) as signal levels increase. The main amplifier may see asame impedance at its output regardless of whether the secondaryamplifiers are idle or fully amplifying, thereby avoiding loadmodulation of the main amplifier that is typical with conventionalDoherty amplifiers. The multiclass no-load-modulation power amplifier'speak efficiency may occur at deeper back-off powers than forconventional Doherty amplifiers.

Some embodiments relate to multiclass power amplifiers. A multiclasspower amplifier may comprise a first amplifier in a first circuit brancharranged to operate in a first amplifier class, a second amplifier in asecond circuit branch arranged to operate in a second amplifier classthat is different from the first amplifier, and an impedance inverterconfigured to receive a combined output from the first amplifier and thesecond amplifier. The multiclass power amplifier may further include athird amplifier in a third circuit branch arranged to operate in thesecond amplifier class and a combining node configured to receive anoutput from the impedance inverter and the third amplifier and provide acombined output to an output port of the multiclass power amplifier fordriving a load.

Some embodiments relate to methods of operating a multiclass poweramplifier. A method may comprise acts of dividing a signal to beamplified into a first signal provided to a first circuit branch, asecond signal provided to a second circuit branch, and a third signalprovided to a third signal branch; amplifying, in the first circuitbranch, the first signal with a first amplifier operated in a firstamplifier class; amplifying, in the second circuit branch, the secondsignal with a second amplifier operated in a second amplifier class thatis different from the first amplifier class; amplifying, in the thirdcircuit branch, the third signal with a third amplifier operated in thesecond amplifier class; combining outputs from the first amplifier andthe second amplifier and providing the combined outputs to an impedanceinverter; combining, at a combining node, an output from the impedanceinverter and an output from the third amplifier; and providing an outputfrom the combining node to an output port of the multiclass poweramplifier.

The foregoing apparatus and method embodiments may be implemented withany suitable combination of aspects, features, and acts described aboveor in further detail below. These and other aspects, embodiments, andfeatures of the present teachings can be more fully understood from thefollowing description in conjunction with the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

The skilled artisan will understand that the figures, described herein,are for illustration purposes only. It is to be understood that in someinstances various aspects of the embodiments may be shown exaggerated orenlarged to facilitate an understanding of the embodiments. The drawingsare not necessarily to scale, emphasis instead being placed uponillustrating the principles of the teachings. In the drawings, likereference characters generally refer to like features, functionallysimilar and/or structurally similar elements throughout the variousfigures. Where the drawings relate to microfabricated circuits, only onedevice and/or circuit may be shown to simplify the drawings. Inpractice, a large number of devices or circuits may be fabricated inparallel across a large area of a substrate or entire substrate.Additionally, a depicted device or circuit may be integrated within alarger circuit.

When referring to the drawings in the following detailed description,spatial references “top,” “bottom,” “upper,” “lower,” “vertical,”“horizontal,” and the like may be used. Such references are used forteaching purposes, and are not intended as absolute references forembodied devices. An embodied device may be oriented spatially in anysuitable manner that may be different from the orientations shown in thedrawings.

FIG. 1 depicts an arrangement of a Doherty amplifier;

FIG. 2A depicts a circuit model for a symmetrical Doherty amplifier whena main amplifier and peaking amplifier are fully amplifying;

FIG. 2B depicts a circuit model for a symmetrical Doherty amplifier whena main amplifier is active and a peaking amplifier is idle(non-amplifying);

FIG. 3 illustrates gain characteristics as a function of output powerfor a Doherty amplifier;

FIG. 4 illustrates efficiency of a Doherty amplifier as a function ofoutput power;

FIG. 5 depicts a no-load-modulation, multiclass power amplifier,according to some embodiments;

FIG. 6A depicts a first circuit model for a no-load-modulation,multiclass power amplifier when all amplifiers are fully amplifying,according to some embodiments;

FIG. 6B depicts a second circuit model for a no-load-modulation,multiclass power amplifier when only a main amplifier is active,according to some embodiments;

FIG. 7 depicts a circuit model for an M-way no-load-modulation,multiclass power amplifier, according to some embodiments; and

FIG. 8 depicts a circuit model for an (N+M+1)-way no-load-modulation,multiclass power amplifier, according to some embodiments.

Features and advantages of the illustrated embodiments will become moreapparent from the detailed description set forth below when taken inconjunction with the drawings.

DETAILED DESCRIPTION

One approach to amplifying signals to high power levels forcommunications is to use a Doherty amplifier, which is depictedschematically in FIG. 1. To aid in understanding the present technology,a summary of Doherty amplification is provided.

A Doherty amplifier 100 may comprise a main power amplifier 132 and apeaking power amplifier 138 that operate in parallel on a signal dividedinto parallel circuit branches. The peaking amplifier 138 is typicallyidle (not amplifying) at low signal levels, and turns on when the mainamplifier 132 begins to saturate. Outputs from the two amplifiers aresubsequently combined into a single RF output.

In further detail, a 90-degree power coupler 110 divides a received RFsignal into two outputs that connect to inputs of the main amplifier 132and the peaking amplifier 138. The power coupler 110 also delays (byapproximately 90 degrees) the phase of the signal provided to thepeaking amplifier with respect to the phase of the signal provided tothe main amplifier. Impedance-matching elements 122, 124 may be placedbefore the main amplifier 132 and peaking amplifier 138. Theseimpedance-matching elements can transform impedances so as to match theinput impedances of the two amplifiers 132, 138 to the impedances of thetransmission lines from the 90-degree coupler 110 or an output impedanceof the coupler 110. Such impedance matching can reduce undesirableeffects of signal reflections from the amplifier inputs.

Additional impedance-matching elements 142, 144 may be located at theoutputs of the main amplifier 132 and peaking amplifier 138 to matchimpedances between the output of the main amplifier 132 to the input ofan impedance inverter 150 (which may be 50 ohms by design) and betweenthe output of the peaking amplifier 138 and an impedance at thecombining node 155 (which may also be 50 ohms). The impedance inverter150 rotates the phase of the signal received from the main amplifier 132by approximately 90 degrees, so that the signals from the main amplifierand peaking amplifier will be essentially in phase at the combining node155. An output impedance-matching element 160 may be used between thecombining node 155 and the Doherty amplifier's RF output to match theoutput impedance of the Doherty amplifier 100 to an impedance of a load(not shown).

In a symmetric Doherty amplifier, the main amplifier 132 and peakingamplifier 138 may be closely similar or identical semiconductor devices.For example, they may be configured to handle a same amount of signalpower and amplify a signal to a same power level when both amplifiersare fully amplifying at their upper limit (e.g., a fully-on state of theDoherty amplifier). Because the input power is split equally to the twoamplifiers, the signal to the main amplifier 132 is typically attenuatedby 3 dB at an output port of the coupler 110 compared to the input RFsignal. Signal values expressed in “dB” refer to relative power levels.

Operational aspects of a Doherty amplifier are illustrated in furtherdetail in FIG. 2A through FIG. 4. FIG. 2A is a simplified circuit modelfor a Doherty amplifier when both the main amplifier 132 and peakingamplifier 138 are fully on (amplifying their respective signals atfull-power values). The main amplifier 132 operating in class AB modeand its output impedance-matching element 142 may be modeled as acurrent source CS_(m) having an internal impedance R and providing nophase delay to the amplified signal. The peaking amplifier 138 operatingin class C mode, its output impedance-matching element 144, and thephase delay of the coupler 110 may be modeled as a second current sourceCS_(p) having an internal impedance R, but providing a 90-degree phasedelay to amplified signals. The impedance inverter 150 may be modeled asa transmission line having a characteristic impedance of R and providinga phase delay of 90 degrees. According to some embodiments, a loaddriven by the Doherty amplifier may have an impedance of R/2.

The phase delays described herein are delays for a carrier wave of aradio-frequency signal that is modulated to encode information onto thecarrier wave. For example, a carrier wave may oscillate at a frequencyhaving a value in a range between 0.7 gigaHertz (GHz) and 7 GHz,depending on what communication protocol is being used (e.g., 2G, 3G,4G, etc.). The main amplifier 132 and peaking amplifier 138 may bedesigned for a particular carrier frequency and communication protocol.As one non-limiting example, an amplifier configured to handle signalsfor 4G communications may be designed for a carrier frequency of 2.6 GHzaccording to one protocol, and the specified phase delays of amplifiercomponents are relative to 2.6 GHz. As another non-limiting example, anamplifier configured to handle signals for 4G communications may bedesigned for a carrier frequency of 1.9 GHz according to anotherprotocol, and the specified phase delays of amplifier components arerelative to 1.9 GHz.

When both the main amplifier 132 and peaking amplifier 138 are activeand driving a load of R/2 with approximately equal amounts of current I(a fully-on state), as depicted in FIG. 2A, straightforward calculationsshow that the main amplifier 132 sees an impedance R at its output, asindicated by the chevron symbol in FIG. 2A. This is referred to as a“1:1 load” condition for the Doherty amplifier. This impedance value canbe calculated in a two-step process. First, the impedance seen lookinginto the combining node 155 from the impedance inverter 150 iscalculated. Second, the impedance looking into the combining node istransformed according to the property of the quarter-wave impedanceinverter 150 to find an impedance (in this case R) looking into theimpedance inverter 150.

FIG. 2B depicts a circuit model for an operating condition of theDoherty amplifier in a fully backed-off state when the peaking amplifier138 is idle (non-amplifying). When an input RF signal to be amplified bythe Doherty amplifier falls below a threshold value, the peakingamplifier 138 becomes idle (non-amplifying) and is modeled essentiallyas an open circuit. For this model, the impedance of the peakingamplifier changes from R to an infinite value in an idle state.Recalculating impedances looking into the combining node 155 and lookinginto the impedance inverter 150 from the main amplifier shows that theimpedance value seen looking into the impedance inverter 150 rises to 2Rin the fully backed-off state. This operating condition is referred toas the “2:1 load” condition of the Doherty amplifier. In this case, themain amplifier's impedance R is no longer well matched to the impedanceit is trying to drive. Such a mismatch can lead to signal reflectionsand inefficient operation of the Doherty amplifier.

The variation in impedance seen by the main amplifier 132 that dependson the state of the peaking amplifier 138 (which is determined by theinput RF signal level) is referred to as “load modulation.” Loadmodulation necessarily adversely affects power-handling capability ofthe amplifier and the amplifier's RF fractional bandwidth. For example,mismatches in impedance cause power reflections, and such reflections tothe main amplifier may constrain the safe operating limit of the mainamplifier appreciably below a power level that it could otherwise handleif there were no power reflections. The amount of reflected power mayfurther depend on frequency, and changes in reflected power withfrequency can take an amplifier out of compliance with a specificationmore quickly (resulting in a narrower bandwidth) than if there were noreflected power.

Additional details of Doherty amplifier gain and efficiency dynamics areillustrated in FIG. 3 and FIG. 4. In FIG. 3, a first gain curve 210(dotted line) depicts gain of a main amplifier 132 as a function ofoutput power P_(out) when the peaking amplifier 138 is idle(non-amplifying). The gain curve 210 corresponds to the 2:1 loadcondition. The peaking amplifier 138 is typically idle at low inputsignal power levels, e.g., input signal levels that will not beginsaturating the main amplifier 132 or signal levels corresponding toabout and more than 6 dB below a peak output power level P_(max) of theDoherty amplifier. These low level signals can be handled by the mainamplifier 132 only. At higher signal levels, the gain of the mainamplifier 132 will begin to saturate and go into “compression,” whichbegins at a power compression point P_(c) and is indicated by thefall-off region 212 in FIG. 3. At this point, the main amplifier 132begins to amplify non-linearly, and would otherwise distort the input RFsignal. The power compression point for a main amplifier 132 will dependupon its design (e.g., the size of active areas in the amplifier'stransistors), and could be any value from 1 Watt to 100 Watts for anamplifier used in a communication system. Smaller or larger values ofthe power compression point may occur in some embodiments.

For a Doherty amplifier, the peaking amplifier 138 begins to amplify theinput RF signal and contribute to the Doherty amplifier's output at thepower compression point P_(c). An example gain curve 230 for the peakingamplifier 138 is also depicted in FIG. 3. The peaking amplifier 138makes up for saturation of the main amplifier 132 at high powers untilthe peaking amplifier begins to saturate, go into compression, and falloff, as indicated in the drawing. Action of the peaking amplifier 138can extend linear amplification by the Doherty amplifier over a range ofhigh powers beyond the capability of the main amplifier 132 alone, untilthe peaking amplifier starts saturating.

FIG. 3 includes a second gain curve 220 for the main amplifier 132 whenthe peaking amplifier 138 is active (amplifying). The curve 220corresponds to the 1:1 load condition. When the peaking amplifier 138 isactive, it effectively adds load impedance to the main amplifier 132(effectively reducing the gain of the main amplifier by about 3 dB) butalso assists in amplifying high power levels (extending the Doherty'scompression to higher powers). FIG. 3 also depicts a gain curve 310(solid dark curve) as a function of output power for the Dohertyamplifier. The Doherty gain curve 310 is a result of the combinedactions of the main amplifier 132 and peaking amplifier 138 as describedabove.

An efficiency curve 410 for a Doherty amplifier is illustrated in FIG.4. The efficiency of the Doherty rises to a peak efficiency E_(p), thatoccurs approximately when the gain of the peaking amplifier 138 hasreached its highest value. Ideally in a Doherty amplifier, the peakefficiency E_(p) would occur at about 6 dB below the maximum outputpower P_(max) (at a power denoted as P_(backoff) in the graph), in aregion referred to as “output power back-off” (OBO, sometimes denotedOPO). The efficiency falls below the peak value E_(p) for output powerlevels below P_(backoff) in a region where the peaking amplifier istransitioning from low gain levels (where the peaking amplifierprimarily loads the main amplifier) to its maximum gain (refer to FIG.3).

In reality, the peak efficiency for a Doherty does not occur at 6 dBOBO, because of several effects present in conventional Dohertyamplifiers. A first effect relates to isolation of the peaking amplifier138 in power back-off. Although the peaking amplifier is modeled aboveas having infinite impedance (open circuit) in back-off, in practicalapplications the impedance is finite at 6 dB OBO. Further, the impedanceinverter 150 and/or output matching elements 142, 144 can exhibit losseswhich may not be insignificant. Additionally, the main amplifier 132 andpeaking amplifier 138 typically have non-ideal I-V curves and/or kneevoltages. All these effects can cause the peak efficiency to occur at avalue that is less than 6 dB OBO (e.g., about 5 dB OBO or less), whichin turn causes the Doherty amplifier's efficiency to be reduced furtherthan shown in FIG. 4 in regions below about 6 dB OBO.

The inventor has recognized and appreciated that load modulation in aDoherty amplifier can adversely affect power handling and bandwidthcapability of a Doherty amplifier. The inventor has also recognized andappreciated that conventional Doherty amplifiers exhibit a peakefficiency in a region between about 5 dB OBO and about 6 dB OBO. Theinventor has further recognized and appreciated that currently-developedcommunication protocols can increase the peak-to-average power ratio(PAPR) in communication signals to 7 dB or more to handle large datarates with high spectral efficiency. As a result, to preserve amplifierlinearity a Doherty amplifier may be operated in a corresponding region(7 dB OBO or more) for a large portion of its operating time, which is aregion where the conventional Doherty amplifier's efficiency isreducing.

The inventor has conceived of a no-load-modulation, improved-efficiency,broadband multiclass power amplifier that can exhibit a peak efficiencyat back-off power margins of 6 dB or more. The amplifier can beessentially free of load modulation effects such as that caused by “on”and “idle” states of the peaking amplifier in a Doherty amplifier. Oneexample of a no-load-modulation, multiclass power amplifier 500 isdepicted in FIG. 5.

A multiclass power amplifier 500 may comprise a plurality of amplifiers532, 536, 537, 538 operating on portions of a received signal (e.g., areceived RF signal) in more than two parallel circuit branches. Thereceived signal at an input port 502 may be divided into the parallelcircuit branches and provided to the plurality of amplifiers. Outputsfrom the amplifiers may be combined at a combining node 155 andsubsequently provided to an output port 580 of the amplifier.

A first amplifier 532 of the plurality of amplifiers may be configuredto operate as a main amplifier in a first amplifying class. For example,the first amplifying class may be class A, class B, or class AB. Theremaining amplifiers 536, 537, 538 may be configured to operate aspeaking amplifiers in a second class (e.g., class C). According to someembodiments, a first portion of the plurality of amplifiers 532, 536 mayoperate on portions of the received signal to be amplified, wherein theportions of the received signal have a first phase. A second portion ofthe plurality of amplifiers 537, 538 may operate on portions of thereceived signal that have a second phase different from the first phase.The second phase may be delayed between 80° and 100° with respect to thefirst phase. In some embodiments, the second phase may be delayedapproximately 90°, or an odd integer multiple thereof, with respect tothe first phase by a coupler 110. In such embodiments, a first combinedsignal from the first portion of the plurality of amplifiers 532, 536may be provided to an impedance inverter 550 that delays the phase ofthe first combined signal with respect to a second combined signal fromthe second portion of the plurality of amplifiers 537, 538 by anapproximately same amount (e.g., between 80° and 100° or approximately90° so that signals combine in phase at the combining node 155.

A no-load modulation, multiclass power amplifier 500 may include one ormore power splitters 510 that divide a received signal into two or moreoutput signals of approximately equal power levels that are provided totwo or more output ports. Outputs from the splitters 510 may be providedto the plurality of amplifiers 532, 536, 537, 538 as depicted in FIG. 5,for example.

In some implementations, the coupler 110 and splitter 510 functionalitymay be combined into a single multi-port device that includes anintegrated network. The integrated network may comprise lumped (e.g.,discrete) and/or distributed (e.g., microstrip waveguides) inductive andcapacitive elements. Examples of lumped inductive elements include, butare not limited to, discrete inductors and bond wires. The multi-portdevice may include at least one input port and four output ports, forexample. The four output ports may provide approximately equal portionsof an input signal received at an input port of the device, although thephase of the signals from two output ports may be delayed byapproximately 90° with respect to the phase of the signals from theother two output ports.

According to some implementations, input impedance-matching elements522, 523, 524, 525 may be located before the inputs to the plurality ofamplifiers 532, 536, 537, 538. An input impedance-matching element 524may comprise lumped and/or distributed inductive and capacitive elementsthat are arranged to match an input impedance of its associatedamplifier 537 to an impedance of an upstream transmission line or outputimpedance from a splitter 510, for example. In some implementations,output impedance-matching elements 542, 543, 544, 545 may be locatedafter outputs from the plurality of amplifiers. An outputimpedance-matching element 544 may comprise lumped and/or distributedinductive and capacitive elements that are arranged to match an outputimpedance of its associated amplifier 537 to an impedance of adownstream transmission line or element, for example.

In some embodiments, the input impedance-matching elements 522, 523,524, 525 and/or output impedance-matching elements 542, 543, 544, 545may be omitted. For example, a single input impedance matching element(not shown) may be located after the input port 502 to match an inputimpedance of the coupler 110 or the above-described multi-port device toan impedance of an upstream transmission line or element. In such acase, the coupler 110 or multi-port device may be constructed to haveoutput impedances that match input impedances of the plurality ofamplifiers. Similarly, an output impedance-matching element 560 andimpedance inverter 550 may be constructed to have an input impedancesthat match to output impedances of the plurality of amplifiers 532, 536,537, 538. The output impedance-matching element 560 may have an outputimpedance that matches to an impedance of a load to be driven by thepower amplifier 500.

A first circuit model 602 for a no-load modulation, multiclass poweramplifier 500 is depicted in FIG. 6A. This simplified model can be usedto represent the power amplifier 500 when all of the plurality ofamplifiers 532, 536, 537, 538 are fully amplifying, each providing apeak output current of I and each having an internal resistance oroutput impedance of R. Two amplifiers may be modeled as two currentsources CS_(p1), CS_(m) connected in parallel and providing amplifiedsignals at a first phase (e.g., approximately 0°. The current sourceCS_(m) may correspond to the main amplifier 532 of the no-loadmodulation, power amplifier 500. The current source CS_(p1) maycorrespond to the first peaking amplifier 536. The amplified signalsfrom the two current sources CS_(p1), CS_(m) may be combined andprovided to an impedance inverter 550. According to some embodiments,the impedance inverter 550 may add a phase delay of an odd multiple of90°, or approximately this value, to the combined signal and provide afirst output signal to a combining node 155. Two additional amplifiers(second and third peaking amplifiers 537, 538) may be modeled as twocurrent sources CS_(p1), CS_(p3) connected in parallel and providingamplified signals at a second phase (e.g., 90° or approximately thisvalue with respect to the first phase) to the combining node 155. Forthe four-amplifier embodiment depicted, a characteristic impedance Z_(c)of the impedance inverter may be approximately R/2. Additionally, animpedance of a load connected to the combining node or an impedance seenat the combining node looking toward the output of the amplifier may beapproximately R/4. In some cases, the impedance R/4 may be an inputimpedance to an output impedance-matching element 560.

Analysis of the circuit model in FIG. 6A, for the impedance values andmatching currents I shown, reveals that an impedance seen by the mainamplifier 532 (modeled as current source CS_(m) in FIG. 6A) is R(indicated by the chevron symbol in the drawing). This impedance valuecan be found in a multi-step analysis. First, an impedance Z_(o) seen atan output of the impedance inverter 550 looking toward the combiningnode 155 is calculated. The value found for that impedance is thenrotated back through the impedance inverter 550 according to theinverter's characteristic impedance to give an impedance Z_(i) seen atthe input of the impedance inverter 550. Finally, the impedance seen bythe main current source CS_(m), which is operating in parallel with oneof the peaking current sources CS_(p1), can be calculated based on theinput impedance Z_(i) at the impedance inverter 550. For the loadimpedance and characteristic impedance shown in FIG. 6A, the impedanceseen by the main current source CS_(m) when all amplifiers (currentsources) are fully amplifying is found to be R.

A circuit model for the four-amplifier embodiment in which only the mainamplifier is active is shown in FIG. 6B. In an ideal model, theimpedances of the amplifiers which are inactive (non-amplifying)increases to an infinite or near infinite value, as indicated in thedrawing. In this case, only the the main amplifier 532 (main currentsource CS_(m)) provides power to the combining node 155, so that theimpedance Z_(o) seen at the output of the impedance inverter is R/4.Rotating this impedance back through the inverter 550, which has acharacteristic impedance Z_(c)=R/2, gives an impedance Z_(i) of R at theinput to the inverter 550. This input impedance is the impedance seen bythe main current source CS_(m) when the peaking current source CS_(p1)is idle (essentially presenting an open circuit).

Based on the analyses for the fully-on state (FIG. 6A) and the fullybacked-off state (FIG. 6B), the main current source CS_(m)(representative of the main amplifier 532) sees a same impedance whenthe peaking current sources CS_(p1), CS_(p2), CS_(p3) are fully activeand when they are idle. A load-modulation effect common to conventionalDoherty amplifiers is not present for the no-load modulation, poweramplifier 500 of the present embodiment.

For the analyses of FIG. 6A and FIG. 6B, the output impedances of theamplifiers (current sources) is taken as R. In practice, the amplifiersmay have an output impedance of a value other than R andimpedance-matching elements 542, 543, 544, 545 may be used to transformthe output impedances to a desired value. In other embodiments, theoutput impedances of the amplifiers may have a value other than R andthe impedance inverter 550 may have a characteristic impedance otherthan R/2, such that it rotates an impedance value at its output to animpedance value that matches an output impedance of the main amplifier532, for example.

FIG. 7 depicts a simplified circuit model 700 for an M-way no-loadmodulation, multiclass power amplifier in which more than fouramplifiers may be arranged in parallel. For the embodiment shown, thereare an even number (M) of amplifiers (represented as current sources inthe circuit model) arranged in parallel. A first half (M/2) of theamplifiers are arranged on a first side of the impedance inverter 750.Of these, one amplifier (CS₁) may be configured to operate as a mainamplifier in class AB mode, for example. The remaining amplifiers (CS₂ .. . CS_(M/2)) on the first side of the impedance inverter 750 may beconfigured to operate as peaking amplifiers (in class C mode, forexample). All of the amplifiers on the first side of the impedanceinverter may be configured to operate on equal portions of a receivedsignal having a first phase (0°, for example).

A second half of the amplifiers (modeled as current sources CS_(M/2+1) .. . CS_(M)) are arranged on a second side of the impedance inverter 750.All of these amplifiers may be configured to operate as peakingamplifiers in a same amplifier class as the peaking amplifiers on thefirst side of the impedance inverter. All of the amplifiers on thesecond side of the impedance inverter 750 may receive equal portions ofthe received signal that are phase delayed (by approximately 90°, forexample) with respect to the phase of signals provided to the first halfof the amplifiers.

The signals from the first half of the amplifiers may be combined andprovided to an impedance inverter 750 having a characteristic impedancevalue of R/(α^(1/2)). A signal from the impedance inverter 750 may beprovided to a combining node 155. Signals from the second half ofamplifiers may be combined and provided to the combining node. Theoutput impedance values for all the amplifiers may be R in someembodiments, though in other embodiments the output impedances may be avalue other than R and impedance-matching elements may be connected tothe outputs of the amplifiers.

Following the analytic process described above in connection with FIG.6A and FIG. 6B, it can be shown that the main amplifier (CS₁) sees aload impedance Z_(off) of approximately R when the multiclass poweramplifier is in a fully back-off state (all peaking amplifiers idle).The analysis also shows that when the power amplifier is fully on, themain amplifier sees a load impedance Z_(on) that is approximately NR/4,where R is an even number greater than 2. When N=4 amplifiers, there isessentially no load modulation in agreement with the 4-way poweramplifier 500 of FIG. 5. When N=6 amplifiers, there may be some loadmodulation, but the load modulation is less than the amount ofmodulation that occurs in a conventional Doherty amplifier. When N=8amplifiers, the load modulation is about equivalent to the loadmodulation that occurs in a conventional Doherty amplifier.

A reduction in load modulation of the main amplifier can improve theperformance of the multiclass power amplifier in several ways. Forexample, a reduction in load modulation can improve the bandwidthperformance of the power amplifier, since the main amplifier sees abetter-matched load whether the power amplifier is in a fully-on stateor fully backed-off state. Additionally, a reduction in load modulationcan reduce the amounts of amplitude-modulation-to-amplitude-modulation(AMAM) distortion and amplitude-modulation-to-phase-modulation (AMPM)distortion that occurs in conventional Doherty amplifiers.

In some embodiments, more than four amplifiers (N>4) may be used in amulticlass power amplifier to improve the amplifier's efficiency. Thelocation of the peak efficiency in back-off, denoted as P_(backoff) inFIG. 3, can be represented approximately by the following equation forthe power amplifier configuration depicted in FIG. 7.P _(backoff) =P _(max)−10 log N  EQ. (1)

In this expresseion, N is the total number of amplifiers in parallelcircuit branches of the multiclass power amplifier. For 4 amplifiers,P_(backoff) occurs at about 6 dB below the amplifier's maximum poweroutput. For 6 amplifiers, P_(backoff) occurs at about 7.8 dB below theamplifier's maximum power output. For 8 amplifiers, P_(backoff) occursat about 9 dB below the amplifier's maximum power output. Multiclasspower amplifier embodiments that include 4, 6, or 8 amplifiers operatingin parallel may be better suited then conventional Doherty amplifiersfor communication systems or applications that havepeak-to-average-power ratios (PAPR) of 7 dB or more.

Additional embodiments of a multi-way, multiclass, no-load-modulationpower amplifier 800 are depicted schematically in FIG. 8. For theseembodiments, there may be an even number or odd number of amplifiers(greater than 2) used in the power amplifier 800. In these embodiments,the amplifiers may provide different amounts of output power or current,so that the main amplifier 810 does not see load modulation,irrespective of the number and arrangement of amplifiers.

The power amplifier 800 may comprise a main amplifier 810 connected on afirst side of an impedance inverter 850. There may also be one or more(N>1) first peaking amplifier(s) (denoted generally with 820) connectedin parallel with the main amplifier on the same side of the impedanceinverter 850. There may be one or more (M>1) second peaking amplifier(s)830 located on a second side of the impedance inverter 850. Outputs fromthe main amplifier 810 and first peaking amplifier(s) 820 may becombined and provided to the impedance inverter 850. Outputs from theimpedance inverter 850 and second peaking amplifier(s) 830 may becombined at a combining node and provided to an output port 880 of thepower amplifier 800, which may connect to a load (shown as having animpedance of R/α). In some implementations, the power amplifier 800 maybe assembled on a circuit board or MMIC.

The main amplifier 810 may be capable of providing a first amount ofmaximum power or current I₁ when fully amplifying a signal to a maximumallowed level. The main amplifier may be configured to operate in afirst amplification class (e.g., class AB). In some embodiments, themain amplifier 810 may comprise one or more semiconductor transistorsformed on a semiconductor die. As just one example, the main amplifier810 may comprise an array of high-electron-mobility transistors (HEMTs)that are formed from gallium-nitride material and located on a firstdie. Other materials and types of transistors may be used in otherembodiments.

The first peaking amplifier(s) 820 may each be capable of providing asecond amount of maximum power or current I₂ when fully amplifying asignal to a maximum allowed level. The first peaking amplifier(s) may beconfigured to operate in a second amplification class (e.g., class C).In some embodiments, the first peaking amplifier(s) 820 may comprise oneor more semiconductor transistors formed on a semiconductor die. As justone example, the first peaking amplifier(s) 820 may comprise an array ofhigh-electron-mobility transistors (HEMTs) that are formed fromgallium-nitride material and located on a second die. Other materialsand types of transistors may be used in other embodiments.

The second peaking amplifier(s) 830 may each be capable of providing athird amount of maximum power or current I₃ when fully amplifying asignal to a maximum allowed level. The second peaking amplifier(s) 830may be configured to operate in the second amplification class (e.g.,class C). In some embodiments, the second peaking amplifier(s) 830 maycomprise one or more semiconductor transistors formed on a semiconductordie. As just one example, the second peaking amplifier(s) 830 maycomprise an array of high-electron-mobility transistors (HEMTs) that areformed from gallium-nitride material and located on a second die. Othermaterials and types of transistors may be used in other embodiments.

In some implementations, the output impedances of the main and peakingamplifiers may be R. In other cases, the output impedance values maydiffer from R, and impedance-matching elements (not shown in FIG. 8) maybe connected to the outputs of one or more amplifiers to match theoutput impedance(s) of the amplifier(s) to downstream impedance values.In other cases, the output impedance values may differ from R, and theoutputs from the impedance inverter and second peaking amplifiers may becombined and provided to an impedance-matching element 860 thattransforms the impedance to match, or approximately match, an impedanceof the load.

Although FIG. 8 does not show how a received signal to be amplified issplit and provided to the different amplifiers, the received signal maybe divided and provided to the amplifiers as depicted in FIG. 5, e.g.,using a combination of a coupler 110 and splitters 510. In embodiments,the received signal to be amplified may be divided such that eachamplifier receives approximately a same input signal level. In someimplementations, signals provided to the second peaking amplifier(s) 830may have a common phase delay (e.g., an odd integer multiple of 90° orapproximately an odd integer multiple of 90° with respect to the commonphase of signals provided to the main amplifier 810 and first peakingamplifier(s) 820. The impedance inverter 850 may provide a compensatingphase delay, such that signals combine at the combining node 155 inphase or approximately in phase.

The multiclass, no-load-modulation power amplifier 800 depicted in FIG.8 may be analyzed in the same way that the amplifiers of FIG. 5 and FIG.7 are analyzed. In the fully backed-off state, the peaking amplifiers820, 830 are idle and are modeled as presenting open circuits (infiniteimpedance). From the model depicted in FIG. 8, it can readily be shownthat the main amplifier 810 sees a load impedance Z_(m,off) of R at itsoutput when the power amplifier is in a fully backed-off state.

When the power amplifier 800 is fully on, then all the peakingamplifiers 820, 830 provide their maximum output I₂, I₃. Under theseconditions, the impedance Z_(o) seen at the output of the impedanceinverter 850 is first found. This value is rotated back through theimpedance inverter 850 based on its characteristic impedance to an inputimpedance value Z_(i). The input impedance value Z_(i) is then used todetermine the impedance Z_(m,on) seen by the main amplifier 810 in thefully-on state. Without being bound to a particular theory, theimpedance Z_(m,on) can be represented by the following expression,according to some embodiments.

$\begin{matrix}{Z_{m,{on}} = {\left\lbrack \frac{\left( {1 + \frac{{NI}_{2}}{I_{1}}} \right)^{2}}{\left( {I_{1} + {NI}_{2} + {MI}_{3}} \right)} \right\rbrack R}} & {{EQ}.\mspace{14mu}(2)}\end{matrix}$Setting the bracketed quotient equal to unity, for no load modulation,gives the following expression.

$\begin{matrix}{I_{3} = {\frac{{NI}_{2}}{M}\left( {1 + \frac{{NI}_{2}}{I_{1}}} \right)}} & {{EQ}.\mspace{14mu}(3)}\end{matrix}$

A power amplifier designer may use EQ. 3 to determine and set a maximumcurrent capability of each amplifier for a no-load-modulation conditiononce the number and arrangement of amplifiers (M and N) has beendetermined. For example, if N=1, M=2, and I₂=I₁, then I₃=I₁, whichrepresents the four-amplifier embodiment depicted in FIG. 5 and analyzedin connection with FIG. 6A and FIG. 6B. In cases where there are an evennumber of amplifiers with one-half of the amplifiers located on one sideof the impedance inverter and the remaining half located on the otherside of the impedance inverter, then M=N+1 and EQ. 3 suggests that anasymmetric configuration of the no-load-modulation power amplifier maybe needed to prevent load modulation. An asymmetric configuration is onein which the total amount of current (or power) provided by amplifierson one side of the impedance inverter 850 does not equal the totalamount of current (or power) provided by amplifiers on the other side ofthe impedance inverter 850 when the no-load-modulation power amplifieris in a fully-on state. EQ. 3 also allows for an odd number ofamplifiers to be used with no load modulation.

According to some embodiments, the location of the peak efficiency atback-off (P_(backoff)) for the power amplifier 800 depicted in FIG. 8may be determined from the following expression.P _(backoff) =P _(max)−10 log(N+M+1)  EQ. (4)The power amplifier 800 embodiments depicted in FIG. 8 may provide moreflexibility in determining the location of the peak efficiency atback-off in addition to providing no load modulation of the mainamplifier 810. Because the main amplifier 810 does not experience loadmodulation, the bandwidth of the power amplifier 800 may be improvedcompared to a Doherty amplifier or other amplifier embodiments in whichthe main amplifier experiences load modulation. Because the location ofP_(backoff) can be pushed to deeper regions of OBO by adding moreamplifiers, the power amplifier 800 can also provide higher operatingefficiency at higher PAPRs.

Methods of operating a no-load-modulation power amplifier of the presentembodiments are also contemplated. A method of operating ano-load-modulation power amplifier of the present embodiments mayinclude a combination of acts such as, but not limited to, receiving, ata coupler, a signal (e.g., an RF signal which may be modulated toinclude data for transmitting), and dividing the signal with the couplerinto a first signal and a second signal. A method may further includesplitting the first signal into a first subset of signals and providingone of the first subset of signals to a main amplifier and the remainderof the first subset of signals to N first peaking amplifiers. A methodmay further include splitting the second signal into a second subset ofsignals and providing the second subset of signals to M second peakingamplifiers. In some cases, a method may include delaying the phase ofthe second signal with respect to the first signal. A method may furtherinclude combining outputs from the main amplifier and the first peakingamplifiers into a first combined output and providing the first combinedoutput to an impedance inverter. A method may also include combiningoutputs from the second peaking amplifiers into a second combined outputand providing the second combined output and an output from theimpedance inverter to a combining node to produce a combined outputsignal that may be provided to an output port of the power amplifier. Insome implementations, the combined output signal may be applied to aload or other apparatus. For example, the combined output signal may beapplied to an antenna to transmit a signal wirelessly. The impedance ofthe load and characteristic impedance of the impedance inverter may havethe relative values described for FIG. 8, for example.

Additional Features for No-Load-Modulation Power Amplifiers

This section describes features that may be included in ano-load-modulation power amplifier of any of the above-describedembodiments.

In some embodiments of power amplifiers depicted in FIG. 5, FIG. 7, andFIG. 8 where the main amplifier and peaking amplifiers are configured tohave a same maximum power or maximum current capability, the mainamplifier and peaking amplifiers may be of a same design. The term “samedesign” means that the amplifiers are nearly identical or identical forall intents and purposes. For example, the amplifiers may be formed asintegrated power transistors (such as the HEMT devices described above)using a same micro fabrication process, though there may be slightvariations in structure due to the nature of the microfabricationprocess. Although the main amplifier and peaking amplifiers may be of asame design, they may be biased differently to operate in differentamplification classes. Where the main amplifier and peaking amplifiersare configured to have different maximum current capabilities, the mainamplifier and peaking amplifiers may comprise semiconductor amplifiersof different designs.

In some embodiments, the main amplifier and peaking amplifiers maycomprise gallium-nitride transistors. In other embodiments, othersemiconductor materials may be used, such as gallium-arsenide orsilicon-germanium, and the invention is not limited to only thesesemiconductors. An example of a semiconductor amplifier that may be usedin whole or in part for the main amplifier and/or peaking amplifiers ina power amplifier of the present embodiments is described in U.S. patentapplication Ser. No. 14/878,952 filed Oct. 8, 2015, titled “TunedSemiconductor Amplifier,” which application is incorporated herein byreference in its entirety, however the invention is not limited to onlythis type of amplifier. In some implementations, the gain values of amain amplifier and a peaking amplifier may be between 16 dB and 30 dB,or approximately between these gain values. In some cases, the gainvalues of the main and peaking amplifiers in a no-load-modulation poweramplifier that is operating in a fully-on state may be the same towithin 3 dB.

As used herein, the phrase “gallium nitride” refers to gallium nitride(GaN) and any of its alloys, such as aluminum gallium nitride(Al_(x)Ga_((1-x))N), indium gallium nitride (In_(y)Ga_((1-y))N),aluminum indium gallium nitride (Al_(x)In_(y)Ga_((1-x-y))N), galliumarsenide phosporide nitride (GaAs_(x)P_(y)N_((1-x-y))N), aluminum indiumgallium arsenide phosporide nitride(Al_(x)In_(y)Ga_((1-x-y))As_(a)Pb_(b) N_((1-a-b))), amongst others. Insome cases, the transistors may be formed from other semiconductormaterials such as gallium arsenide, silicon carbide, silicon germanium,silicon, indium phosphide, etc. and the invention is not limited togallium-nitride-based amplifiers.

Because a no-load-modulation power amplifier of the present embodimentsincludes more than two semiconductor amplifiers, the power-handlingcapability of each semiconductor amplifier is reduced compared to thepower-handling capability of a conventional Doherty amplifier in whichonly two semiconductor amplifiers are used (for a fixed powerspecification). For example, in a case where a power amplifier 500 ofthe present embodiments includes four amplifiers that are configured tohandle a same amount of power at maximum output, each amplifier onlyneeds to provide one-half the amount of power compared to atwo-amplifier Doherty. Additionally, since each amplifier has aboutone-half the power-handling requirement the amplifiers can be madesmaller and exhibit less drain-to-source parasitic capacitance. Thecomparative reduction in parasitic capacitance can allow each amplifierto operate at higher speeds and support larger RF fractional bandwidthsand video bandwidths.

A no-load-modulation power amplifier of the present embodiments may beimplemented in a variety of packages. According to some embodiments, ano-load-modulation power amplifier may be assembled on a printed-circuitboard or application board using discrete components. In someembodiments, a no-load-modulation power amplifier may be fabricated aspart of a monolithic microwave integrated circuit (MMIC). For example,the main and peaking amplifiers may each be fabricated as one or moreintegrated, semiconductor transistors that are mounted on a PCB or MMIC.The coupler, delay elements, and impedance-matching elements may beformed as discrete or integrated components, or a combination thereof.Connections between components may be made using wire bonds, microstriptransmission lines, or a combination thereof. Discrete, lumped, orintegrated components (e.g., capacitors, diodes, inductors, etc.) may beconnected in a power amplifier circuit.

A no-load-modulation power amplifier of the present embodiments may beincorporated into a cell phone or base station amplification system andused for amplifying wireless communication signals, according to someembodiments. A no-load-modulation power amplifier of the presentembodiments may be incorporated into any device that has wireless accesscapability including, but not limited to, computers, tablets, smartphones, smart watches, vehicles, smart appliances, etc. Advantages of ano-load-modulation power amplifier of the present embodiments forportable devices include its higher bandwidth and higher gaincapabilities compared to conventional Doherty amplifiers. The increasedefficiency can contribute to prolonged battery life in a portable deviceand better accommodate signals with higher PAPRs. A broader bandwidthcan cause less signal distortion compared to a conventional Dohertyamplifier and accommodate higher data rates.

Referring again to FIG. 5, the inventor has recognized and appreciatedthat removing the impedance-matching elements 542, 543, 544, 545 beforethe impedance inverter 550 can be beneficial. In such cases, the signalsfrom the main and peaking amplifiers may be combined first and animpedance-matching element may be added after the combining node totransform an impedance after the combining node to match, orapproximately match, the load impedance. The inventor has found fromsimulations that the impedance-matching element before and after theamplifiers may add electrical path length that cannot be correctlycompensated for by an impedance inverter 550 that provides only a 90°delay. Accordingly, the delay at the impedance inverter may be requiredto be an odd integer multiple of 90° greater than 1. Simulations showthat such an increase in the delay at the impedance inverter canundesirable narrow the RF fractional bandwidth of the power amplifier.By removing the impedance-matching elements 542, 543, 544, 545, thedelay at the impedance inverter may be 90° or approximately 90°, whichimproves the RF fractional bandwidth and video bandwidth performance ofthe power amplifier.

In some implementations, the impedance inverter of a no-load-modulationpower amplifier of the present embodiments may be implemented in wholeor in part as a conductive strip line (e.g., a microstrip transmissionline) that has a length L and width W. In some implementations, theremay be bond wires that connect drain pads of the main and peakingamplifiers to the strip line. The length L of the conductive strip linemay be between approximately 2 millimeters and approximately 6millimeters, according to some embodiments, and may be selected toprovide a desired inductance for the strip line. The width of theconductive strip line may be between approximately 100 microns andapproximately 1000 microns, according to some embodiments, and may beselected to provide a desired inductance for the strip line. Byselecting the length and width of the strip line, the distributedinductance of the strip line 810 may be tuned to a desired value.According to some embodiments, a total of the distributed inductance ofthe strip line may be between approximately 250 picoHenries andapproximately 1.5 nanoHenries.

In some implementations, the conductive strip line is formed over aground conductor or ground plane and separated from the ground conductoror ground plane by a dielectric material (not shown). In otherembodiments, the conductive strip line may not be formed over oradjacent to a ground plane. Instead, a ground plane may be removed froman area of a PCB at which the conductive strip line is patterned. Theconductive strip line, when implemented in the impedance inverter for RFsignals, may comprise an integrated, distributed impedance element whichis essentially entirely inductive. In some implementations, the stripline may include some parasitic capacitance and resistance. An examplesubstrate on which a conductive strip line is formed may comprise aprinted circuit board in some embodiments, a high-frequency laminatecapable of carrying signals at GHz frequencies in some embodiments, aceramic, or a semiconductor. An example of a high-frequency laminate islaminate model RO4003® available from Rogers Corporation of Chandler,Ariz.

In some embodiments, an impedance inverter of a no-load modulation poweramplifier of the present embodiments may not include a microstrip line.Instead, an impedance inverter may be embodied as an RF networkcomprising bond wires and capacitive elements. The bond wires may becomprise gold or any other suitable conductor and may have a diameterbetween 20 microns and 80 microns. The spacing between the bond wiresthat connect to drain pads on a main amplifier or peaking amplifier maybe between approximately 100 microns and approximately 800 microns,according to some embodiments. The bond wires may comprise lumpedinductive elements of the impedance inverter. Such bond wires arerecognized in the field of RF electronics as “lumped inductors” havingan inductance that is determined primarily by a length and diameter ofthe bond wire.

In some implementations, a combining node 155 of a no-load modulationpower amplifier of the present embodiments may be located at one or moredrain pads of the peaking amplifier(s) that are located on a same sideof the impedance inverter as the load.

According to some embodiments, a value of load impedance R seen by themain amplifier in a no-load modulation power amplifier of the presentembodiments may be set to a value that is based upon the design of themain amplifier. For example, a main amplifier may be rated at a maximumdrain-to-source current I_(max) for an applied operating voltage V_(ds)that is applied between the amplifier's drain and source. The resistanceR at which maximum power may be transferred from the main amplifier maybe determined approximately from the following relation.R≈2(V _(ds) −V _(k))/I _(max)  (EQ. 5)where V_(k) is the knee voltage for the amplifier.

In view of the foregoing, various embodiments of no-load-modulationpower amplifier and methods of operating a no-load-modulation poweramplifier may be implemented as follows.

(1) A multiclass power amplifier embodiment comprising: a firstamplifier in a first circuit branch arranged to operate in a firstamplifier class; a second amplifier in a second circuit branch arrangedto operate in a second amplifier class that is different from the firstamplifier; an impedance inverter configured to receive a combined outputfrom the first amplifier and the second amplifier; a third amplifier ina third circuit branch arranged to operate in the second amplifierclass; and a combining node configured to receive an output from theimpedance inverter and the third amplifier and provide a combined outputto an output port of the multiclass power amplifier for driving a load.

(2) The multiclass power amplifier of embodiment (1), wherein the firstamplifier sees a same impedance at its output when the second amplifierand the third amplifier are idle and when the second amplifier and thethird amplifier are fully amplifying.

(3) The multiclass power amplifier of embodiments (1) or (2), whereinthe first amplifier operates in class AB mode and the second amplifierand the third amplifier operate in class C mode.

(4) The multiclass power amplifier of any one of embodiments (1) through(3), wherein the load has an impedance of approximately R/α and acharacteristic impedance of the impedance inverter is approximatelyR/(α½).

(5) The multiclass power amplifier of any one of embodiments (1) through(4), wherein the value of a is 4.

(6) The multiclass power amplifier of any one of embodiments (1) through(5), wherein the impedance inverter is configured to provide a phasedelay at a carrier-wave frequency that is approximately equal to an oddmultiple of 90 degrees.

(7) The multiclass power amplifier of any one of embodiments (1) through(6), wherein the impedance inverter comprises a microstrip transmissionline.

(8) The multiclass power amplifier of any one of embodiments (1) through(7), wherein R is an impedance seen by the first amplifier and isapproximately equal to an impedance value at which a maximum amount ofpower is transferred from the first amplifier.

(9) The multiclass power amplifier of any one of embodiments (1) through(8), wherein a peak efficiency for the multiclass power amplifier occursbetween 6 dB and 12 dB output power back-off.

(10) The multiclass power amplifier of any one of embodiments (1)through (9), further comprising an impedance-matching element connectedbetween the combining node and an output of the multiclass poweramplifier.

(11) The multiclass power amplifier of any one of embodiments (1)through (10), further comprising a fourth amplifier arranged to operatein the second amplifier class having an output that is coupled to thecombining node, wherein, when fully amplifying, average power levels ofthe second amplifier, the third amplifier, and the fourth amplifier areapproximately equal to an average power level of the first amplifier.

(12) The multiclass power amplifier of any one of embodiments (1)through (11), further comprising: a first impedance-matching elementconnected to an input of the first amplifier; a secondimpedance-matching element connected to an input of the secondamplifier; and a third impedance-matching element connected to an inputof the third amplifier.

(13) The multiclass power amplifier of any one of embodiments (1)through (12) incorporated in a wireless communication apparatus.

Any of the amplifier embodiments (1) through (13) may be used with anyof the following method embodiments (14) through (22).

(14) A method of operating a multiclass power amplifier, the methodcomprising: dividing a signal to be amplified into a first signalprovided to a first circuit branch, a second signal provided to a secondcircuit branch, and a third signal provided to a third signal branch;amplifying, in the first circuit branch, the first signal with a firstamplifier operated in a first amplifier class; amplifying, in the secondcircuit branch, the second signal with a second amplifier operated in asecond amplifier class that is different from the first amplifier class;amplifying, in the third circuit branch, the third signal with a thirdamplifier operated in the second amplifier class; combining outputs fromthe first amplifier and the second amplifier and providing the combinedoutputs to an impedance inverter; combining, at a combining node, anoutput from the impedance inverter and an output from the thirdamplifier; and providing an output from the combining node to an outputport of the multiclass power amplifier.

(15) The method of embodiment (14), further comprising presentingapproximately a same impedance to the first amplifier irrespective ofwhether the second amplifier and third amplifier are fully amplifying orare idle.

(16) The method of embodiment (14) or (15), further comprisingexhibiting a peak efficiency for the power amplifier that occurs between6 dB and 12 dB output power back-off.

(17) The method of any one of embodiments (14) through (16), furthercomprising delaying, by the impedance inverter, a phase of a carrierwave passing through the impedance inverter by a value that isapproximately equal to an odd multiple of 90 degrees.

(18) The method of any one of embodiments (14) through (17), furthercomprising transforming, with an impedance-matching element, animpedance between the combining node and an output of the poweramplifier.

(19) The method of any one of embodiments (14) through (18), whereinamplifying, in the first circuit branch, the first signal comprisesoperating the first amplifier in class AB mode.

(20) The method of any one of embodiments (14) through (19), whereinamplifying, in the second circuit branch, the second signal comprisesoperating the second amplifier in class C mode.

(21) The method of any one of embodiments (14) through (20), furthercomprising amplifying, with the power amplifier, communication signalsin a cell phone base station.

(22) The method of any one of embodiments (14) through (21), furthercomprising amplifying, with the power amplifier, a communication signalfor a cell phone or wireless communication device.

CONCLUSION

The terms “approximately” and “about” may be used to mean within ±20% ofa target value in some embodiments, within ±10% of a target value insome embodiments, within ±5% of a target value in some embodiments, andyet within ±2% of a target value in some embodiments. The terms“approximately” and “about” may include the target value.

The technology described herein may be embodied as a method, of which atleast some acts have been described. The acts performed as part of themethod may be ordered in any suitable way. Accordingly, embodiments maybe constructed in which acts are performed in an order different thandescribed, which may include performing some acts simultaneously, eventhough described as sequential acts in illustrative embodiments.Additionally, a method may include more acts than those described, insome embodiments, and fewer acts than those described in otherembodiments.

Having thus described at least one illustrative embodiment of theinvention, various alterations, modifications, and improvements willreadily occur to those skilled in the art. Such alterations,modifications, and improvements are intended to be within the spirit andscope of the invention. Accordingly, the foregoing description is by wayof example only and is not intended as limiting. The invention islimited only as defined in the following claims and the equivalentsthereto.

What is claimed is:
 1. A multiclass power amplifier comprising: a main amplifier in a first circuit branch arranged to operate in a first amplifier class; at least one first peaking amplifier in a second circuit branch arranged to operate in a second amplifier class that is different from the first amplifier class, the second circuit branch being in parallel with the first circuit branch; an impedance inverter configured to receive a combined output from the main amplifier and the at least one first peaking amplifier; a plurality of second peaking amplifiers arranged to operate in the second amplifier class; and a combining node configured to receive an output from the impedance inverter and a combined output from the plurality of second peaking amplifiers and provide a combined output to an output port of the multiclass power amplifier for driving a load.
 2. The multiclass power amplifier of claim 1, wherein the main amplifier sees a same impedance at its output when the at least one first peaking amplifier and the plurality of second peaking amplifiers are idle and when the at least one first peaking amplifier and the plurality of second peaking amplifiers are fully amplifying.
 3. The multiclass power amplifier of claim 1, wherein the main amplifier operates in class AB mode and the at least one first peaking amplifier and the plurality of second peaking amplifiers operate in class C mode.
 4. The multiclass power amplifier of claim 1, wherein the load has an impedance of approximately R/a and a characteristic impedance of the impedance inverter is approximately R/(α^(1/2)).
 5. The multiclass power amplifier of claim 4, wherein the value of α is
 4. 6. The multiclass power amplifier of claim 1, wherein the impedance inverter is configured to provide a phase delay at a carrier-wave frequency that is approximately equal to an odd multiple of 90 degrees.
 7. The multiclass power amplifier of claim 1, wherein the impedance inverter comprises a microstrip transmission line.
 8. The multiclass power amplifier of claim 4, wherein R is an impedance seen by the main amplifier and is approximately equal to an impedance value at which a maximum amount of power is transferred from the main amplifier.
 9. The multiclass power amplifier of claim 1, wherein a peak efficiency for the multiclass power amplifier occurs between 6 dB and 12 dB output power back-off.
 10. The multiclass power amplifier of claim 1, further comprising an impedance-matching element connected between the combining node and an output of the multiclass power amplifier.
 11. The multiclass power amplifier of claim 1, wherein, when fully amplifying, average power levels of the at least one first peaking amplifier and plurality of second peaking amplifiers are approximately equal to an average power level of the main amplifier.
 12. The multiclass power amplifier of claim 1, further comprising: a first impedance-matching element connected to an input of the main amplifier; and a second impedance-matching element connected to an input of the at least one first peaking amplifier.
 13. The multiclass power amplifier of claim 1, incorporated in a wireless communication apparatus.
 14. A method of operating a multiclass power amplifier, the method comprising: dividing a signal to be amplified into a first signal provided to a first circuit branch, a second signal provided to a second circuit branch, and a third signal provided to a third circuit branch; amplifying, in the first circuit branch, the first signal with a main amplifier operated in a first amplifier class; amplifying, in the second circuit branch, the second signal with at least one first peaking amplifier operated in a second amplifier class that is different from the first amplifier class, the second circuit branch being in parallel with the first circuit branch; amplifying, in the third circuit branch, the third signal with a plurality of second peaking amplifiers operated in the second amplifier class; combining outputs from the main amplifier and the at least one first peaking amplifier and providing the combined outputs to an impedance inverter; combining, at a combining node, an output from the impedance inverter and a combined output from the plurality of second peaking amplifiers; and providing an output from the combining node to an output port of the multiclass power amplifier.
 15. The method of claim 14, further comprising presenting approximately a same impedance to the main amplifier irrespective of whether the at least one first peaking amplifier and the plurality of second peaking amplifiers are fully amplifying or are idle.
 16. The method of claim 14, further comprising exhibiting a peak efficiency for the power amplifier that occurs between 6 dB and 12 dB output power back-off.
 17. The method of claim 14, further comprising delaying, by the impedance inverter, a phase of a carrier wave passing through the impedance inverter by a value that is approximately equal to an odd multiple of 90 degrees.
 18. The method of claim 14, further comprising transforming, with an impedance-matching element, an impedance between the combining node and an output of the power amplifier.
 19. The method of claim 14, wherein amplifying, in the first circuit branch, the first signal comprises operating the main amplifier in class AB mode.
 20. The method of claim 14, wherein amplifying, in the second circuit branch, the second signal comprises operating the at least one first peaking amplifier in class C mode.
 21. The method of claim 14, further comprising amplifying, with the power amplifier, communication signals in a cell phone base station.
 22. The method of claim 14, further comprising amplifying, with the power amplifier, a communication signal for a cell phone or wireless communication device. 